Lightweight Vivaldi Antenna for High-Voltage Ultra-Wideband Systems

1. Introduction

Ultra-wideband (UWB) antennas are key enabling components of modern wireless technologies that require high speed [1], high bandwidth[2], diversity [3,4], and low signal distortion [5]. Applications range from high-speed data transmission [6], water detection [7], landmine detection [8], to biomedical imaging [9,10], radar [11], through-wall radar (TWR) [12], fluid properties determination [13], and intentional electromagnetic interference [14], as shown in Figure 1. One of the most widely used antennas for UWB applications is the Vivaldi antenna, a type of tapered slot antenna (TSA). Originally called Vivaldi aerial [15], the Vivaldi antenna is characterized by its high gain, wide bandwidth, low cross-polarization level, and stable radiation characteristics. Various optimization techniques have been proposed to enhance its performance, including feeding mechanisms, slot integration, radiator shape modifications, inclusion of dielectric lenses, parasitic patches between the two radiator arms, edge corrugations on the arms, and the use of metamaterials [16].

Vivaldi antennas are commonly used in ground-penetrating radar (GPR) due to the characteristics mentioned above and also due to their compactness [17]. GPR antennas must meet the following conditions: they must be broadband, have narrow beam-width radiation patterns, and be capable of radiating medium-power pulses. In the case of bi-static GPR, low crosstalk is also required. Crosstalk can be reduced by using arrays with directive antennas, such as Vivaldi [18,19] and shielded bowtie [20] antennas. Besides, it is necessary to consider the near-field interaction with the soil and the target, which affects crosstalk, impedance matching, and detection efficiency. Stadler et al. [21] show that full-wave simulation approach including the antenna model is required to accurately determine the near-field antenna coupling with the soil, which has frequency dependent permittivity in the GPR operational frequency range [22].

To achieve broadband performance, geometries such as the hybrid antenna presented in [23] can be used. This design combines two modes of operation: a broadband monopole mode between 200 MHz and 400 MHz, and a Vivaldi antenna mode between 400 MHz and 25 GHz. In this case, the radiation pattern is not critical, as the antenna is designed to be used in a reverberation chamber (RC), where mode stirrers generate an isotropic average antenna pattern.

UWB antennas are also used to characterize radiated pulsed emanations of secure communications systems [24]. Typical spectral content of radiated signals from components used in quantum communications, for instance, ranges from 40 MHz to 1 GHz [25]. UWB directive antennas, such as double-ridge guide horn, are preferred for electromagnetic (EM) emission measurements. However, to achieve low frequency operation, size and weight scale up resulting in large (i.e. squared meter aperture or cross-section) and heavy (i.e. tens of kg) antennas [26].

UWB antennas for electromagnetic compatibility (EMC) tests must have stable characteristics with frequency. In [27], the development of a Vivaldi antenna for electromagnetic field measurements in EMC tests between 0.5 GHz and 4 GHz is presented. This antenna is a low-cost alternative compared to other options covering the same frequency range and allows the replacement of multiple narrowband antennas traditionally used for these tests.

Directive antennas are preferred for radiated immunity and intentional electromagnetic interference (IEMI) testing. These tests require compliance with field uniformity criteria on surfaces where the devices under test (DUT) are located. A hyperband antenna for IEMI applications is presented in [28]. In addition to achieving the field uniformity criterion, this antenna achieves the radiation of high-power pulses with amplitudes on the order of 6 kV. The bandwidth of this antenna is also wide, as it can radiate pulses with a rise time on the order of 90 ps. Recently, tests have been reported on this antenna with source voltages up to 24 kV [29].

One option to achieve radiation patterns with high gain and beam-steering capability is through the use of antenna arrays. An example of such an array, with broadband operation, is presented in [30]. This array is capable of radiating high power for use in high-power jamming systems. To prevent arc flashes when handling high power, the antenna tips are rounded to reduce the electric field intensity around them. Additionally, the array is constructed entirely of metal, as antennas manufactured on substrates tend to have low durability when operated at high power. An 8x8 array with a ±45° slant configuration operates between 2 GHz and 6 GHz, achieving gains between 18 dBi and 21 dBi for various beam-steering angles.

In [31], a linear 1x10 array of antipodal Vivaldi antennas is presented, where the effect of adding metallic inserts to the metallizations and slots of each element is explored. The best performance is observed when two inserts are added to the slots, resulting in an operating frequency range from 5.7 GHz to 12.0 GHz, a gain between 12 dBi and 19 dBi within this range, and a radiated power between 0.75 W and 0.9 W.

In [32], a compact Vivaldi antenna array for UWB pulse radiation is proposed. To compact the array, the ends of the radiating elements are bent in both directions. To expand the impedance bandwidth, short-circuited branches are inserted between the bent radiating elements and the ground plane. This design results in an 8x4 array capable of radiating UWB pulses with an impedance bandwidth ranging from 0.6 GHz to 6 GHz, and a compact size of 2.5 cm x 2.5 cm x 4.5 cm. The gain varies between 5 dBi and 20 dBi.

Antennas capable of radiating pulses must be analyzed not only in the frequency domain but also in the time domain. While parameters such as gain, reflection coefficient, and group delay as a function of frequency are important, they are insufficient for this type of antenna. It is necessary to characterize the distortion introduced by the antenna on the transmitted pulses [33]. This time-domain characterization includes parameters such as pulse width extension and fidelity factor to establish the similarity between the transmitted and received pulses [34].

A common application of antennas capable of radiating high-power pulses is through-wall radar and imaging of nearby objects. For through-wall radar (TWR), antennas with high gain, wide bandwidth, and effective wall penetration capabilities are required. These features are provided by the Vivaldi antenna, which also exhibits directional radiation patterns that enhance wall penetration. In [35], structural optimizations, feeding techniques, and performance improvement strategies for Vivaldi antennas are presented to enhance their imaging and detection capabilities for through-wall radar applications.

For imaging nearby objects, [36] describes the design of an antipodal Vivaldi antenna (AVA) capable of radiating impulse-type waveforms. The antenna operates between 1.8 GHz and 10 GHz, and its design focuses on optimizing its pulse response in the time domain. This involves analyzing the transfer function between two identical antennas, the shape of the received pulse, group delay, and gain. The antenna successfully reconstructs two radar targets positioned 7 cm apart.

A comparison of the relevant parameters for various UWB antennas for EMC, IEMI, and radar applications is presented in Table 1. The table shows that the starting operation frequency of the UWB antennas is higher than 0.5 GHz for most of the cases, even though the applications may require to cover lower frequencies. This is due to size and weight constraints. To overcome this problem, in this paper, we propose a lightweight UWB differential antenna with a starting frequency of 0.33 GHz suitable for GPR, EMC, and EMI applications. The antenna presented here is optimized for wide-band impedance matching up to 4.2 GHz. The frequency domain and also the time domain performances are analyzed in detail.

This paper is organized as follows. Section II outlines the antenna design to achieve the desired impedance matching. Section III describes the procedure for implementing and testing the antenna, presenting the obtained reflection coefficient and the radiated electric field. Section IV details the characterization of the designed antenna in the time domain, presenting an analysis of the waveform responses and calculating the pulse width stretch ratio (SR) for different pulses over the main planes of the antenna.

Figure 2 summarizes the workflow followed in this study: (i) geometry definition and Ansys HFSS-based parametric design of a differential Vivaldi with exponential inner/outer tapers and a PLA spacer, using lightweight aluminum–polyester–aluminum sheets; (ii) prototype fabrication and high-voltage feed implementation with ferrite-ring and tapered-coax baluns, qualified to 12.4 kV; (iii) frequency-domain validation with a VNA showing a measured impedance bandwidth of 335.8 MHz–4.2 GHz; and (iv) time-domain assessment using a 110 ps input pulse and an $S_{21}$-derived transfer function to compute the radiated field $E_r\left(t\right)$, followed by pulse-fidelity analysis—pulse-width stretch ratio $<10\%$ up to $\pm {30}^{\circ }$—and a $\sim {30}^{\circ }$ 3 dB beamwidth for UWB pulses. This process map clarifies how the lightweight, wideband, and high-voltage requirements were achieved and verified.

2. Antenna Design

Various design approaches have been proposed to improve the performance of Vivaldi antennas in ground-penetrating radar (GPR) applications. These include the incorporation of combined slots at the edges to extend the electrical length and achieve frequencies from 75 MHz [38], the optimization of feeding strategies such as microstrip, curved microstrip, and CPW to expand bandwidth and gain [39], the inclusion of elliptical slots in antipodal E-shaped configurations to reduce the cutoff frequency [40], and the design of S-band Vivaldi arrays optimized by parametric sweeps to improve coupling and directivity [41].

Various antenna configurations have been proposed to improve the performance of GPR systems. These include a suspended elliptical patch with an optimized slot that achieves bandwidths of 878–1260 MHz and gain >8 dBi [42]; a low-profile bowtie antenna with a loaded elliptical patch and reflector plane, which reduces the minimum frequency to 0.18 GHz and achieves compact dimensions with 7.41 dBi gain [43]; and a slotted bowtie with bent arms and absorbent backing, covering 0.5–3 GHz and offering >30 dB isolation [44].

UWB antennas such as the one designed in this article can also be used in applications that require high-capacity, low-latency and reliable communications in 5G/6G networks [45], as well as in satellite links using MIMO-UWB systems, which increase spectral efficiency and support emerging applications such as IoT and massive communications [46].

In electromagnetic compatibility tests, antennas are made to withstand high voltages through the use of multilayer lenses that adjust dielectric constants and improve directivity [47], and geometric optimization of the aperture and propagation in TEM horn antennas [48].

Three main goals are addressed in the design using novel optimization and fabrication techniques: (i) wide-band impedance matching by optimizing the geometry and materials through parametric simulations, (ii) light weight by using aluminum composite boards, and (iii) high voltage by isolating the differential feed point.

2.1. Geometry and Materials

The configuration of the proposed Vivaldi antenna is shown in Figure 3. This antenna is designed from two identical conductive arms (radiating arms) with an insulator of polylactic acid (PLA) between the arms. The material of the radiating arms is aluminum. The antenna size is set at 64.2 ($x_1$) x 48 ($y_1$) cm².

The inner curves of the antenna (see Figure 3) follow an exponential function described by:

$$y=F_1{e}^{Mx}+F_2$$

(1)

where,

$$F_1={\displaystyle \frac{(y_d-y_b)}{(e^{M{x}_c}-e^{M{x}_a})}}$$

(2)

$$F_2={\displaystyle \frac{(y_b{e}^{M{x}_a}-y_d{e}^{M{x}_c})}{(e^{M{x}_c}-e^{M{x}_a})}}$$

(3)

The outer curves of the antenna (see Figure 3) follow an exponential function described by:

$$y_D=E_1{e}^{Nx}+E_2$$

(4)

where,

$$E_1={\displaystyle \frac{(y_h-y_f)}{(e^{N{x}_g}-e^{N{x}_e})}}$$

(5)

$$E_2={\displaystyle \frac{(y_f{e}^{N{x}_e}-y_h{e}^{N{x}_g})}{(e^{N{x}_g}-e^{N{x}_e})}}$$

(6)

The equations symbols are presented in Table 2 and Figure 3. The values of these symbols are given in Table 3.

Equations 1 - 6 were used to obtain an initial state for the antenna. Parametric analysis was then used in Ansys HFSS to maximize bandwidth. This analysis included optimization of the outer curve for low frequency performance, optimization of the inner curve for high frequency performance, and optimization of arm separation for impedance matching.

2.2. Weight Reduction

To reduce the overall weight of the antenna, each conductive arm was replaced by a layered structure with one dielectric layer of 2 mm in between two conductive layers of 0.5 mm. That is, from bottom to top there is a conductive layer (aluminum), then a second dielectric layer (polyester) and finally a third conductive layer (aluminum). Using aluminum composite panels provides a weight reduction of 66.7% when compared to same-size aluminum sheet. Figure 4 shows the final configuration of the proposed Vivaldi antenna. The simulation was conducted using Ansys HFSS, where the dimensions of the radiation box were adjusted to 14.5 x 14.5 x 14.5 cm³.

Table 3 shows the parameters and their values for the proposed antenna. Figure 5 shows the $S_{11}$ parameter of the proposed antenna with and without insulator (i.e. PLA). It is observed that the antenna matching improves with the presence of the insulator located between the conductive arms.

2.3. Feed Point

The antenna feed point was designed to handle high voltage and provide a wideband transition to a single-ended coaxial transmission line. In addition to improve the matching impedance, the PLA strip between the two arms also provides electric insulation. It was experimentally verified that the design has a breakdown voltage of 12.4 kV. Figure 6 shows a spark produced during the test when the high voltage applied at the feed point exceeds this threshold. A balun was implemented using two ferrite rings of 4W620 material from Würth Elektronik on the differential transmission line at the feed point. This provides common-mode current rejection up to 1 GHz. To provide a smooth transition for higher frequencies, a coaxial cable tapered balun was implemented at the connection point by gradually removing the screen of the coaxial cable.

2.4. Gain

The maximum gain of the main lobe of the designed Vivaldi antenna was simulated in Ansys HFSS, with the results shown in Figure 7. The gain varies between -7.66 dBi and 13.88 dBi over a frequency range of 0.1 GHz to 3 GHz. The gain is positive throughout the frequency range from 0.15 GHz to 3 GHz, except for a negative gain point of -1.09 dBi at 0.65 GHz. This negative gain point coincides with a region of antenna mismatch (see Figure 11). The antenna exhibits peak gain values at frequencies of 0.25, 0.5, 0.95, 2.3, and 3 GHz, achieving gains of 6.72, 7.61, 6.24, 10.87, and 13.88 dBi, respectively.

2.5. Radiation Patterns

The normalized gain radiation patterns of the antenna are plotted in the E-plane (xy-plane in Figure 3) and H-plane (yz-plane in Figure 3), as shown in Figure 8 and Figure 9, respectively, at frequencies distributed throughout the antenna’s matching range. The radiation patterns show uniformity in their shape in the two main planes, although as the frequency increases, more secondary lobes appear. The radiation patterns presented confirm the uniformity of the maximum gain as a function of frequency.

3. Antenna Testing

The proposed Vivaldi antenna was fabricated and measured as shown in Figure 10. The arms were implemented with a sheet composed by two layers of aluminium filled with polyester composite. The layered sheet provides the required strength for the arms size and light weight. Total weight of the antenna is less than 600 g. A 3D printed PLA insulator between the arms was added to adjust the matching impedance. The insulator also enables the antenna to operate at high voltages as discussed before.

The measurement was conducted using a portable network vector analyzer. Figure 11 shows the measured reflection coefficient of the Vivaldi antenna. The impedance bandwidth obtained from the measurements is 335.8 MHz to 4.2 GHz. A slight mismatch is presented at 600 MHz. This small discrepancy between simulated and measured results are due to the inclusion of elements added during the fabrication, which were neglected in the simulation, such as the ferrites and the wooden support.

This impedance bandwidth corresponds to a fractional bandwidth of 1.70, classifying the antenna as an ultra-wideband (UWB) device for radar and communication applications, and as a hyperband (HB) device for electromagnetic interference applications [49].

3.1. Radiated Electric Field Measurement

The electric field radiated by the Vivaldi antenna was measured using a reference biconical antenna as shown Figure 12. Both antennas were $h=1.5$ m high and separated by a distance $d=2.9$ m. The S-parameters between the reference antenna and the antenna under test were measured as presented in the diagram of Figure 13 to obtain the transfer function between the incident voltage wave at port 2, $b_2$, and the reflected wave at port 1, $a_1$. That relationship, shown in (7), is the voltage forward gain (VFG). In our measurement, VFG is the ratio between the received voltage, $V_r$, and the voltage applied to the Vivaldi antenna, $V_s$, in the frequency range where both antennas are impedance matched.

$$S_{21}={\displaystyle \frac{b_2}{a_1}}$$

(7)

To calculate the radiated electric field $E_r$, the antenna factor of the receiving antenna $AF$ can be used as shown in (8).

$$AF={\displaystyle \frac{E_r}{V_r}}$$

(8)

Using (7), (8), and replacing $a_1$ and $b_2$ by $V_s$ and $V_r$, respectively, the radiated electric field can be calculated as

$$E_r=AF·S_{21}·V_s$$

(9)

Figure 14 shows the electric field radiated by the Vivaldi antenna scaled to 1 m in vertical polarization. 6 measurements were conducted and normalized to 1 m. The figure shows a wide-band frequency response with a resonant transfer function between 0.15 and 3 GHz.

4. Time-Domain Characteristics

4.1. Waveform Responses

To obtain the received electric field $E_r$ for an input signal corresponding to a wideband pulse, in (9) $V_s$ is replaced by the pulse shown in Figure 15, whose rise time is 110 ps.

The frequency spectrum obtained for $E_r$ is converted to the time domain using the inverse Fourier transform, resulting in the measured pulse presented in Figure 16. The received pulse corresponds to the electric field scaled to 1 m. The figure shows the comparison of the calculated signal and the simulated pulse from Ansys HFSS, obtained as the parameter $rE$.

In Figure 16 good agreement is observed in the shape and width of the simulated and measured pulses with small differences in the amplitude and width of the two positive peaks. This figure shows that the radiated electric field waveform is the derivative of the applied voltage typically produced by the Vivaldi antenna [14,50].

To determine the similarity between the received and the input pulses, they are compared in the frequency domain (Figure 17). The frequency spectra are obtained through the Fourier transform of the pulses. A comparison of the spectra shows that they are centered around the same frequency (1.875 GHz). Despite minor amplitude differences, the spectra satisfactorily include signals within the range of 0.5 GHz to 3.5 GHz.

4.2. Pulse Width Stretch Ratio (SR)

To analyze how the received pulses stretch compared to the input pulse, the parameter pulse width stretch ratio (SR) is used, as defined in (10) and (11) [33,51].

$$E_s\left(t\right)={\displaystyle \frac{{\int }_{-\infty }^t{\left|s\left(t^{\prime }\right)\right|}^{2}d{t}^{\prime }}{{\int }_{-\infty }^{\infty }{\left|s\left(t^{\prime }\right)\right|}^{2}d{t}^{\prime }}}$$

(10)

$$SR={\displaystyle \frac{E_{s2}^{-1}\left(0.95\right)-E_{s2}^{-1}\left(0.05\right)}{E_{s1}^{-1}\left(0.95\right)-E_{s1}^{-1}\left(0.05\right)}}$$

(11)

In the numerator of (11), the width of the pulse of interest (signal $s_2$) is calculated as the time interval between the times where the accumulated pulse energy reaches 5% and 95% of its total energy. This width is compared to that of a reference signal (signal $s_1$), calculated in the same manner, in the denominator of (11).

In this case, the pulses of interest, $s_2$, correspond to the electric field scaled to a distance of 1 m on the main lobe of the antenna, in both the E-plane and H-plane, at angles of 0° (maximum radiation direction), 15°, 30°, -15°, and -30°. These pulses are shown in Figure 18 and Figure 19.

Two pulses were used as reference signal $s_1$ in (11): the input voltage $V_s$, shown in Figure 15, and its derivative. The derivative was used because, as previously mentioned, in a pulse-radiating antenna, the radiated electric field has the form of the derivative of the input voltage [14,50]. Since the pulses of interest $s_2$ are radiated electric fields, it is appropriate to use the derivative of the voltage as a reference. The input voltage itself is also used as a reference to evaluate the stretching of the radiated pulses relative to the driving signal. The pulse width stretch ratio (SR) calculated for the two reference signals $s_1$ is presented in Table 4 and Table 5.

As expected, the SR value is closer to 1.0 when the derivative of the input voltage is used as the reference. For both reference signals, the SR values increase as the direction moves away from the maximum radiation. Using the input voltage as the reference, the stretching ranges from 33.50% to 40.56%, while using its derivative as the reference, the stretching ranges only from 3.86% to 9.36%. This demonstrates the antenna’s good performance in preserving the input pulse width, by keeping the stretching below 10% when using a reference signal with the same shape as the radiated electric field.

The results in time domain show that the proposed antenna is suitable to transmit/receive radiated pulses with rise time as low as 110 ps maintaining the waveform. Also, it was shown that the pulse-width SR is lower than 10% up to 30° in aperture. These properties are highly desirable for pulsed sensing applications, such as GPR or WPR, since they determine the resolution in space and the capacity of discriminating contiguous objects. The directionality of the antenna provides better performance against interference in the measurements produced by objects located behind the antenna. Directionality is desirable for sensing and also for EMI testing. The 3dB beamwidth obtained for a pulsed signal, as shown in Figure 18, is 30°.

5. Conclusions

The design and verification process for an ultra-wideband high-voltage Vivaldi antenna is presented. The antenna is implemented using a sheet composed of two aluminum layers filled with a polyester composite to reduce its weight. The operational frequency range of the designed antenna was optimized through a parametric variation on the arms geometry from 335.8 MHz to 4.2 GHz, enabling it to radiate pulses with rise time of about 110 ps. With the PLA insulator inserted between the arms of the antenna to form an insulated-coplanar-plates transmission line, it can operate at voltages of up to 12.4 kV and can provide broadband impedance matching to a standard 50 Ohm system. This capability makes the designed Vivaldi antenna suitable for applications in radar, electromagnetic compatibility (EMC), or intentional electromagnetic interference (IEMI). The antenna maintains the radiated pulse width very close to that of the driving pulse, achieving a maximum pulse width stretch ratio (SR) of 1.0936 for variations from $-{30}^{\circ }$ to ${30}^{\circ }$ in the E-plane and H-plane. This result translates into a low level of distortion in the radiated pulse. It was shown also that the 3 dB beamwidth is 30° for a UWB pulsed signal, which is desirable for sensing and EMI testing.