Design and Implementation of a Novel DC-DC Converter with Step-Up Ratio and Soft-Switching Technology

1. Introduction

In recent years, renewable energy has gained increasing attention, with photovoltaic power generation system becoming one of the most widely applied forms of renewable energy. These systems are characterized by sustainable energy supply and minimal environmental impact. With advancements in technology, the installation and operational costs of photovoltaic power generation systems have continuously decreased, and their long-term maintenance costs are also relatively low, making them an ideal energy supply option. Additionally, with the rapid development of power electronics technology, converters play a crucial role in photovoltaic power generation systems, as they can significantly improve the energy conversion efficiency of the system. The primary function of a converter is to convert the direct current (DC) generated by the photovoltaic module array into alternating current (AC) or other DC values, for use in the power grid or energy storage. Due to the relatively low voltage output of photovoltaic modules and the need for subsequent applications in high-voltage systems, boost converters [1,2,3] are commonly used in photovoltaic power generation systems. However, boost converters also have certain drawbacks. When the switch is operated within a reasonable duty cycle, in order to prevent overheating and damage to the switch during prolonged operation, the voltage gain of the converter is limited, thus failing to meet the requirement for high output voltage. Moreover, if the switching frequency is too high when operating the switch, it results in significant switching losses. Although soft switching technologies [4,5,6,7] can reduce switching losses, they increase the size and cost of the converter and still cannot achieve the required higher output voltage. These issues affect the overall performance and cost-effectiveness of the photovoltaic power generation system.

In recent years, many researchers have begun to explore high step-up converters [8,9,10,11] to address the drawbacks of traditional boost converters. High step-up converters can convert the relatively low output voltage from a photovoltaic module array to a higher voltage, thereby reducing energy losses and improving system efficiency. However, this technology also faces several challenges. The design of high step-up converters is more complex and requires advanced electronic components and control techniques, which leads to increased manufacturing costs and greater risks to system reliability. Additionally, high step-up converters operate in high-voltage environments, posing potential voltage breakdown and insulation issues. Therefore, special insulation designs and protection measures are necessary. The boost converter presented in reference [8] demonstrates excellent step-up performance. However, due to the use of a three-winding coupled inductor, which increases the voltage gain through the turns ratio, the converter's physical size becomes significantly larger. In addition, the main switch in this converter does not achieve soft switching, resulting in higher switching losses during transitions between the on and off states. Furthermore, as the output voltage increases, the main switch must withstand higher voltage and current levels, significantly reducing the switch's lifespan. Reference [9] proposes a multi-stage series connection to increase the voltage gain; but the conversion efficiency decreases as the number of stages increases. Reference [10] presents an inductively coupled soft switching bidirectional converter, which offers a high step-up ratio and reduces input voltage ripple. However, it requires five switch components and six capacitors, which increases the size and cost of the physical hardware. Reference [11] introduces an interleaved high voltage ratio boost converter with coupled inductor. However, during the interleaved operation, the inactive converter still transmits some power, which reduces the overall efficiency and causes distortion in the average current. Reference [12] proposed an inductively coupled soft-switching bidirectional converter that also achieves a high step-up ratio. However, its main switch components must withstand higher voltage switching stresses, requiring switches with higher voltage ratings.

To address the drawbacks of the aforementioned converters, this paper proposes a high step-up soft switching converter. This converter uses a coupled inductor to replace the conventional energy storage inductor and incorporates a boost circuit, enabling a higher voltage conversion ratio under the same duty cycle, thereby improving the voltage gain of the converter. In addition, this paper incorporates a resonance branch into the converter and uses a simple switch signal control to achieve ZVS of the main switch, reducing switching losses in hard switching high step-up converters and improving conversion efficiency. First, the circuit architecture of the proposed converter was established using PSIM simulation software [13], and its feasibility was verified through simulation analysis. Finally, the implementation was carried out using the TMS320F2809 digital signal processor [14] from Texas Instruments, which demonstrated the superior conversion performance of the high step-up soft-switching converter.

2. The Proposed High Step-Up Converter

2.1. Operating Principle of the High Step-Up Hard Switching Converter

Figure 1 depicts the circuit architecture of the proposed high step-up hard-switching converter. This design replaces the traditional energy storage inductor with a coupled inductor and incorporates an additional boost circuit formed by $D_1$ and $C_1$. By leveraging the turns ratio of the coupled inductor and the boost circuit, the voltage conversion ratio is significantly enhanced. Furthermore, the converter features a simple circuit structure and ease of control, making it advantageous for practical applications. The operation of the converter can be divided into two modes based on the state of the switch (on and off). When the switch is on, the duty cycle $D$ of the converter during one period T is defined by Equation (1).

$$D\triangleq {\displaystyle \frac{t_{\mathrm{on}}}{T}}={\displaystyle \frac{t_{\mathrm{on}}}{t_{\mathrm{on}}+t_{\mathrm{off}}}}$$

(1)

Where $t_{\mathrm{on}}$ is the time during which the switch is conducting within one period, and $t_{\mathrm{off}}$ is the time during which the switch is off.

(1) 

Switch on ($0\le t_{\mathrm{on}}\le DT$)

When the switch S is conducting, the diode $D_2$ is also conducting, while the diode $D$ and $D_1$ are in the off state. The equivalent circuit is shown in Figure 2, and the turns ratio N of the coupled inductor is defined by Equation (2). At this point, the inductor voltages $v_{L1}$ and $v_{L2}$ are expressed by Equations (3) and (4), respectively, while the voltage $v_{C2}$ across the energy storage capacitor $C_2$, is given by Equation (5).

$$N\stackrel{\Delta }{=}{\displaystyle \frac{N_2}{N_1}}$$

(2)

$$v_{L1}=V_i$$

(3)

$$v_{L2}={\displaystyle \frac{N_2}{N_1}}{V}_i=N{V}_i$$

(4)

$$v_{C2}=v_{L2}+v_{C1}=V_i\left(N+{\displaystyle \frac{1}{1-D}}\right)$$

(5)

(2) 

Switchoff ($DT\le t_{\mathrm{off}}\le T$)

When the switch S is off, diode $D_2$ is in the off state, while diodes $D$ and $D_1$ are conducting. The equivalent circuit is shown in Figure 3, and the inductor voltages $v_{L1}$ and $v_{L2}$ are given by Equations (6) and (7), respectively.

$$v_{L1}=V_i-v_{C1}=-V_i{\displaystyle \frac{D}{1-D}}$$

(6)

$$v_{L2}=V_i+v_{C2}-v_{L1}-V_o=N{V}_i+2{\displaystyle \frac{V_i}{1-D}}-V_o$$

(7)

Based on the volt-second balance theorem for the inductor $L_2$, the Equations (4) and (7) lead to Equation (8). After simplification, the voltage conversion ratio between the output voltage $V_o$and the input voltage $V_i$ can be expressed by Equation (9).

$$N{V}_{i}DT+(N{V}_i+2{\displaystyle \frac{V_i}{1-D}}-V_o)(1-D)T=0$$

(8)

$$G={\displaystyle \frac{V_o}{V_i}}={\displaystyle \frac{2+N}{1-D}}$$

(9)

From Equation (9), the relationship between the voltage gain of the converter and the duty cycle is shown in Table 1. From Table 1 and Figure 4, it can be observed that, for the same duty cycle, the voltage conversion ratio of the converter can be increased by adjusting the turns ratio of the coupled inductor.

2.2. Operating Principle of the High Step-Up Hard Switching Converter

The high step-up hard switching converter described above can increase the converter’s voltage conversion ratio by utilizing a coupled inductor and an additional boost circuit. However, if the voltage or current cannot be reduced to zero before switching occurs, switching losses will arise, which in turn lowers the overall efficiency of the converter. To improve the conversion efficiency, this paper proposes a high step-up soft switching converter with an inductive coupling. The circuit architecture of this converter is shown in Figure 5, while Figure 6 illustrates the switching control signals for the converter [15]. The converter achieves a high voltage conversion ratio through inductive coupling and an additional boost circuit. It also incorporates a resonant branch composed of $L_r$, $C_1$ and $S_r$, where $C_1$ is the capacitor originally used in the hard switching converter circuit. By controlling the switching signals, the converter enables ZVS for the main switch, thus improving efficiency and reducing switching losses. This paper will analyze the operating modes of the proposed soft switching converter, which is divided into nine distinct modes. Figure 7 shows the switching waveforms for each component. Before explaining the operating modes, the following assumptions are made:

1) The converter operates in continuous conduction mode (CCM), and the circuit has reached a steady state.

2) All components are ideal, meaning that during conduction they are treated as short circuits, and during cutoff, as open circuits. Consequently, there is no conduction voltage drop across the power switching components.

3) The input and output voltages are maintained at constant values.

4) The currents of the energy storage inductors $L_1$ and $L_2$ are considered constant (i.e. $i_{L1}=I_{L1}$, $i_{L2}=I_{L2}$).

(1) 

Mode 1($t_0~t_1$)

When operating in Mode 1, the equivalent circuit is shown in Figure 8. In this mode, the main switch S is in the off state, while the auxiliary switch S_r is turned on first. As a result, the resonant capacitor $C_1$ begins to discharge, and the voltage across the resonant inductor $L_r$ is $v_{C1}$. The current through the resonant inductor $i_{Lr}$ rises from zero. Therefore, the resonant inductor $L_r$ and resonant capacitor $C_1$ form a resonant tank, and the circuit equations can be expressed by Equation (10). After solving, the $i_{Lr}$ and $v_{C1}$ are given by Equation (11), while the resonant impedance $Z_o$ and resonant angular frequency ${\omega }_o$ are given by Equation (12). When the voltage $v_{C1}$ across the resonant capacitor drops to $V_i$, diode $D_1$ switches from the off state to the on state. The circuit then transitions to Mode 2. The time for this transition is given by Equation (13).

$$\left\{\begin{array}{l}{i}_{Lr}(t_o)=0\\ v_{C1}(t_o)={\displaystyle \frac{V_i}{1-D}}\\ i_{C1}(t)=-i_{Lr}(t)\\ v_{C1}(t)=v_{Lr}(t)\end{array}\right.\text{，}{t}_0\le t\le t_1$$

(10)

$$\left\{\begin{array}{l}{i}_{Lr}(t)={\displaystyle \frac{V_i}{(1-D)Z_o}}\sin {\omega }_o(t-t_o)\\ v_{C1}(t)={\displaystyle \frac{V_i}{1-D}}\cos {\omega }_o(t-t_o)\end{array}\right.\text{，}{t}_0\le t\le t_1$$

(11)

Where the resonance impedance $Z_o\triangleq \sqrt{{\displaystyle \frac{L_r}{C_1}}}$ and resonance frequency

$${\omega }_o\triangleq {\displaystyle \frac{1}{\sqrt{L_r{C}_1}}}$$

(12)

$$T_1=t_1-t_o=\sqrt{L_r{C}_1}{\cos }^{-1}\left(1-D\right)$$

(13)

(2) 

Mode2 ($t_1~t_2$)

When entering working Mode 2, the equivalent circuit is shown in Figure 9. At this point, the voltage across the resonant capacitor $v_{C1}$ has decreased to $V_i$. At this point, the auxiliary switch S_r remains on, and both the resonant inductor $L_r$ and the resonant capacitor $C_1$ continue to form a resonant circuit. The resonant inductor current $i_{Lr}$ will continue to increase, and the resonant capacitor $C_1$ will keep discharging. The circuit equations for this mode can be expressed as Equation (14). From Equation (14), $i_{Lr}(t)$ and $v_{C1}(t)$ can be derived as shown in Equation (15). When the voltage across the resonant $v_{C1}$ capacitor drops to zero, at which point ${\omega }_o(t-t_1)={\displaystyle \frac{\pi }{2}}$, the resonant inductor current $i_{Lr}$ can be calculated from Equation (15) as shown in Equation (16). The operating time for this mode can be derived as shown in Equation (17). At this point, the anti-parallel diode of the main switch S will begin to conduct, and the system will enter working Mode 3.

$$\left\{\begin{array}{l}{i}_{Lr}(t)+C_1{\displaystyle \frac{d{v}_{C1}(t)}{dt}}=I_{L1}\hfill \\ \begin{array}{l}{L}_r{\displaystyle \frac{d{i}_{Lr}(t)}{dt}}=v_{C1}(t)\\ v_{C1}(t_1)=V_i\end{array}\hfill \end{array}\right.\text{，}{t}_1\le t\le t_2$$

(14)

$$\left\{\begin{array}{l}{i}_L{­}_r(t)=I_{L1}+{\displaystyle \frac{V_i}{Z_o}}\sin {\omega }_o(t-t_1)\\ v_{C1}(t)=V_i\cos {\omega }_o(t-t_1)\end{array}\right.\text{，}{t}_1\le t\le t_2$$

(15)

$$i_L{­}_r(t_2)=i_S{­}_r(t_2)=I_{L1}+{\displaystyle \frac{V_i}{Z_o}}$$

(16)

$$T_2=t_2-t_1={\displaystyle \frac{\pi }{2}}\sqrt{L_r{C}_1}$$

(17)

(3) 

Mode 3 ($t_2~t_3$)

When the system enters working Mode 3, the equivalent circuit is shown in Figure 10. At this point, the voltage across the resonant capacitor $v_{C1}$ has dropped to a very small negative voltage, which causes the anti-parallel diode of the main switch S to forward conduct. As a result, the voltage across the main switch S is zero. The control of the main switch S is then switched from the off state to the on state, achieving ZVS for the main switch S. At the same time, the auxiliary switch S_r is switched off. The circuit equations for this mode can be expressed as Equation (18). To ensure ZVS for the main switch, the delay time $t_d$ must satisfy the condition in Equation (19), and typically $t_d$ is 5% to 10% of the switching period T. Moreover, to ensure that the main switch S can still achieve ZVS under heavy load conditions, and considering the turn-off speed of the auxiliary switch S_r, an additional time delay $t_{\alpha }$ must be considered. Thus, the operating time $t_{DSr}$ of the auxiliary switch can be expressed as shown in Equation (20).

$$\left\{\begin{array}{l}{i}_S(t_2)=I_{L1}-i_L{­}_r(t_2)=-{\displaystyle \frac{V_i}{Z_o}}\\ i_L{­}_r(t_2)=I_{L1}+{\displaystyle \frac{V_i}{Z_o}}\end{array}\right.\text{，}{t}_2\le t\le t_3$$

(18)

$$t_d\ge T_1+T_2=\sqrt{L_r{C}_1}\left({\displaystyle \frac{\pi }{2}}+{\cos }^{-1}\left(1-D\right)\right)$$

(19)

$$t_{DSr}\ge t_d+t_{\alpha }=\sqrt{L_r{C}_1}\left({\displaystyle \frac{\pi }{2}}+{\cos }^{-1}\left(1-D\right)\right)+t_{\alpha }$$

(20)

(4) 

Mode4 ($t_3~t_4$)

In Mode 4, the main switch S is conducting, while the auxiliary switch S_r switches off. Diodes $D_1$ and $D_2$ are forward-biased, and the equivalent circuit is shown in Figure 11. In this mode, to allow the energy storage capacitor $C_2$ and the resonant inductor $L_r$ to form a resonant tank, the capacitance of $C_2$ is chosen to be close to that of $C_1$. Additionally, the resonant inductor $L_r$ and the common-mode inductance $L_2$ both discharge into the energy storage capacitor$C_2$. At this point, the current through the resonant inductor $L_r$ is $I_{L1}+{\displaystyle \frac{V_i}{Z_o}}$. The circuit equation for this mode can be expressed as Equation (21). Solving for $i_{Lr}(t)$ and $v_{C2}(t)$ gives Equation (22). In this working mode, the current through the resonant inductor $i_{Lr}$ decreases from $I_{L1}+{\displaystyle \frac{V_i}{Z_o}}$ to zero, and diodes $D_1$ and $D_2$ switch off. The circuit then transitions into Mode 5.

$$\left\{\begin{array}{l}{L}_r{\displaystyle \frac{d{i}_{Lr}\left(t\right)}{dt}}=v_{L2}(t)-v_{C2}(t)\hfill \\ C_2{\displaystyle \frac{d{v}_{C2}\left(t\right)}{dt}}=i_{Lr}(t)\hfill \\ v_{L2}(t)=N{v}_{L1}(t)=N{V}_i\hfill \\ v_{C1}\left(t_3\right)=0\hfill \\ i_{Lr}(t_4)=0\hfill \end{array}\right.\text{，}{t}_3\le t\le t_4$$

(21)

$$\left\{\begin{array}{l}{i}_{Lr}(t)=\left(I_{L1}+{\displaystyle \frac{V_i}{Z_o}}\right)\cos {\omega }_o(t-t_3)+{\displaystyle \frac{N{V}_i}{Z_o}}\sin {\omega }_o(t-t_3)\\ v_{C2}(t)=N{V}_i\left[1+\cos {\omega }_o(t-t_3)\right]+\left(I_{L1}+{\displaystyle \frac{V_i}{Z_o}}\right)\sin {\omega }_o(t-t_3)\end{array}\right.$$

(22)

(5) 

Mode5 ($t_4~t_5$)

In Mode 5, the equivalent circuit is shown in Figure 12. At this point, the main switch S remains conducting. The circuit equation for this mode can be expressed as Equation (23). This mode continues until the main switch S transitions from conducting to off.

$$\left\{\begin{array}{l}{i}_S(t)=I_{L1}\hfill \\ v_{C1}(t)=0\hfill \end{array}\right.\text{，}{t}_4\le t\le t_5$$

(23)

(6) 

Mode6 ($t_5~t_6$)

In Mode 6, the main switch S is turned off and the equivalent circuit is shown in Figure 13. As a result, the current $I_{L1}$ charges the resonant capacitor $C_1$, causing the voltage across the resonant capacitor$v_{C1}$ to gradually increase. The circuit equation for this mode is given by Equation (24). By solving Equation (24), $v_{C1}(t)$ can be obtained as Equation (25). When the resonant capacitor voltage $v_{C1}$ reaches $V_i$, this mode ends and transitions into Mode 7. Therefore, the operating time for this mode can be determined from Equation (26).

$$C_1{\displaystyle \frac{d{v}_{C1}(t)}{dt}}=I_{L1}\text{，}{t}_5\le t\le t_6$$

(24)

$$v_{C1}(t)={\displaystyle \frac{I_{L1}}{C_1}}(t-t_5)$$

(25)

$$T_6=t_6-t_5={\displaystyle \frac{V_i{C}_1}{I_{L1}}}$$

(26)

(7) 

Mode7 ($t_6~t_7$)

In Mode 7, the voltage across the resonant capacitor $v_{C1}$ is $V_i$, and the main switch S remains in the off state, while the auxiliary switch S_r is also off. As a result, the coupled inductor $L_2$ and the energy storage capacitor $C_2$ transfer energy to the load via diode D. The circuit equation for this mode is given by Equation (27), and the equivalent circuit is shown in Figure 14. By solving Equation (27), the voltage $v_{C1}$ across the resonant capacitor $C_1$ can be obtained as Equation (28). At this point, the current flowing through the coupled inductor $I_{L1}$ and the current $I_{L2}$ gradually decrease, and when the currents $I_{L1}$ and $I_{L2}$ become equal, the circuit transitions into Mode 8.

$$\left\{\begin{array}{l}{C}_1{\displaystyle \frac{\mathrm{d}{v}_{C1}}{\mathrm{d}t}}=I_{L1}-I_{L2}\\ v_{C1}(t_6)=V_i\end{array}\right.\text{，}{t}_6\le t\le t_7$$

(27)

$$v_{C1}(t)={\displaystyle \frac{I_{L1}-I_{L2}}{C_1}}(t-t_6)\text{，}{t}_6\le t\le t_7$$

(28)

(8) 

Mode8 ($t_7~t_8$)

In Mode 8, the voltage across the resonant capacitor $v_{C1}$ is $V_i/(1-D)$, and diode $D_1$ transitions from the conducting to the off state. The resonant capacitor $C_1$ stops charging, and both the main switch S and the auxiliary switch S_r remain off. Consequently, the coupled inductor and the energy storage capacitor $C_2$ continue to transfer energy to the load. The equivalent circuit for this mode is shown in Figure 15, and the circuit equation is given by Equation (29). When the voltage across the energy storage capacitor $C_2$ drops to zero, this mode ends, and the circuit transitions into Mode 9.

$$\left\{\begin{array}{l}{I}_{L1}=I_{L2}\\ v_{C1}(t_7)={\displaystyle \frac{V_i}{1-D}}\end{array}\right.\text{，}{t}_6\le t\le t_7$$

(29)

(9) 

Mode9 ($t_8~t_9$)

In Mode 9, the voltage across the energy storage capacitor $v_{C2}$ is zero, and diode D transitions from conducting to the off state. Both the main switch S and the auxiliary switch S_r remain off. The equivalent circuit for this mode is shown in Figure 16, marking the completion of the entire switching cycle analysis. The cycle will repeat once the auxiliary switch S_r turns on again, returning to Mode 1 of the next cycle.

To demonstrate the performance of the proposed high step-up soft switching converter, a comparison is made with various high step-up soft switching converters. The comparison is based on voltage gain, switch voltage ratings, number of switching components, number of diode components, number of inductive components, and number of capacitive components. The results are summarized in Table 2. As shown in the table, the proposed converter achieves a high voltage gain with fewer components and utilizes simple signal control for soft switching, highlighting its key advantages.

3. Component Design of the Proposed High Step-Up Converter

The output power of the high step-up hard switching converter proposed in this paper is 340W. The relevant circuit parameters and specifications are provided in Table 3.

In the case where all components are operating under ideal conditions, the input power should be equal to the output power, i.e.

$$V_i{I}_{L1}={\displaystyle \frac{V_o^2}{R}}$$

(30)

$$I_{L1}={\displaystyle \frac{V_o^2}{V_i^2}}{\displaystyle \frac{V_i}{R}}$$

(31)

Substituting Equation (9) into Equation (31) yields, as expressed in Equation (32).

$$I_{L1}={\left({\displaystyle \frac{2+N}{1-D}}\right)}^2{\displaystyle \frac{V_i}{R}}$$

(32)

where ${\displaystyle \frac{2+N}{1-D}}$ represents the voltage gain of the converter.

When the main switch S is turned on, $v_{L1}$ can be expressed as:

$$v_{L1}=V_i=L_1{\displaystyle \frac{\mathrm{d}{i}_{L1}}{\mathrm{d}t}}$$

(33)

From Equation (33), it can be observed that when the main switch is turned on, the inductor current $i_{L1}$ increases linearly, with the conduction time denoted as $t_{\mathrm{on}}=DT$. The rate of change of the inductor current can be calculated as shown in Equation (34).

$$\Delta i_{L1}{­}_{(\mathrm{close})}={\displaystyle \frac{V_i}{L_1}}DT$$

(34)

Using Equations (32) and (34), the maximum and minimum values of the inductor current $i_{L1}$ can be derived, as expressed in Equations (35) and (36) [16].

$$I_{L1(\max )}=I_{L1}+{\displaystyle \frac{\Delta i_{L1}}{2}}={\left({\displaystyle \frac{2+N}{1-D}}\right)}^2{\displaystyle \frac{V_i}{R}}+{\displaystyle \frac{V_{i}D}{2L_{1}f}}$$

(35)

$$I_{L1(\min )}=I_{L1}-{\displaystyle \frac{\Delta i_{L1}}{2}}={\left({\displaystyle \frac{2+N}{1-D}}\right)}^2{\displaystyle \frac{V_i}{R}}-{\displaystyle \frac{V_{i}D}{2L_{1}f}}$$

(36)

If $I_{L1(\min )}=0$, the inductor can operate at the boundary between CCM and discontinuous conduction mode, which yields:

$${\left({\displaystyle \frac{2+N}{1-D}}\right)}^2{\displaystyle \frac{V_i}{R}}={\displaystyle \frac{V_{i}D}{2L_{1(\min )}f}}$$

(37)

From Equation (37), the value of $L_{1(\min )}$ can be calculated as:

$$L_{1(\min )}={\displaystyle \frac{DR}{2f}}{\left({\displaystyle \frac{1-D}{2+N}}\right)}^2$$

(38)

Since the coupled inductor structure is similar to the coupled transformer structure, the maximum value of $I_{L2}$ can be shown as [16]:

$$I_{L2(\max )}=-{\displaystyle \frac{I_{L1(\max )}}{1+N}}$$

(39)

3.1. Design of Coupled Inductors

The given scenario involves ensuring that a converter operates in CCM across all duty cycles, with a maximum output power of 340W, a fixed output voltage of 430V, and a rated load of 550Ω. From the relationship between the duty cycle D and the ${\displaystyle \frac{D}{2}}{\left({\displaystyle \frac{1-D}{2+N}}\right)}^2$ function curve shown in Figure 17, it is observed that the function ${\displaystyle \frac{D}{2}}{\left({\displaystyle \frac{1-D}{2+N}}\right)}^2$ reaches its maximum value when D=0.33. By substituting the load resistance R=550Ω, the turns ratio of the coupled inductors N=2, the switching frequency f=25kHz, and the duty cycle D=0.33 into Equation (38), the minimum primary inductance $L_{1(\min )}$ of the coupled inductor is calculated to be 102μH. To ensure that the coupling inductor $L_1$ operates in CCM under both light and heavy load conditions, the calculated inductance value is further multiplied by a safety factor of 1.25. Hence, the selected value of the coupling inductor $L_1$ is 127 μH.

3.2. Design of Capacitors $C_1$ and $C_2$

From Figure 1 of the high step-up hard switching converter, it can be observed that the capacitor $C_1$, coupled inductor $L_1$, main switch S, and diode $D_1$ form a traditional boost converter. When the duty cycle of the main switch S is 0.8, the voltage across capacitor $C_1$ is approximately 360V. From Equation (5), under the same operating conditions, the voltage across capacitor $C_2$ is around 500V. Therefore, the rated voltage of capacitor $C_1$ is selected as 400V, while the rated voltage of capacitor $C_2$ is 600V. Based on the analysis of operating Mode 3 in Equation (18) and Figure 7, it is evident that the capacitance value of $C_1$ influences the reverse current flowing through the main switch S. Considering ripple size and component availability, the chosen capacitance value for $C_1$ is 0.33μF/400V. Additionally, from the explanation of operating Mode 4 of this high step-up soft-switching converter, it is understood that $C_2$ should have a capacitance value similar to that of $C_1$, so $C_2$ is selected as 0.33μF/600V.

3.3. Design of Resonant Inductor $L_r$

Since the conduction time of the auxiliary switch in a typical soft switching converter is usually designed to be between 5% and 10% of the switching period, let $t_d=10\%T=4\mu \mathrm{s}$ ($T={\displaystyle \frac{1}{f}}={\displaystyle \frac{1}{25\mathrm{kHz}}}=40\mu \mathrm{s}$) and $t_{\alpha }=2\%T=0.8\mu \mathrm{s}$ represent these values, respectively. From Equation (20), if the duty cycle D is set within the range of 0.1 to 0.8, the resonance inductance value can be calculated to vary between 8.08μH and 17.08μH. Based on this, the chosen resonance inductance for the converter is 18μH.

After the design of the aforementioned components, the specifications of the components used in the proposed high step-up soft-switching converter are listed in Table 4.

4. Simulation Results

The PSIM simulation software [13] was first used in this study to model the high step-up soft switching converter, in order to verify that the proposed converter achieves superior conversion performance under both light and full load conditions. Figure 18 shows the waveform of the duty cycle, input voltage, and output voltage when the converter operates at full load (P=340W). The output voltage of 430V is obtained when the input voltage is 72V and the duty cycle is 0.33, as shown in Figure 18. Therefore, the converter exhibits a high step-up ratio, and its voltage gain is consistent with the value listed in Table 1. Figure 19 and Figure 20 show the simulation waveforms of the electrical quantities for the main switch S and auxiliary switch S_r when the converter operates at full load (P=340W). Figure 21 and Figure 22 show the simulation waveforms of the electrical quantities for the main switch S and auxiliary switch S_r when the converter operates at light load (P=100W). From Figure 19 and Figure 21, it can be observed that at both full load (P=340W) and light load (P=100W), the voltage across the main switch S drops to zero before the switch enters the conduction mode. This demonstrates that the main switch S achieves ZVS, confirming that the proposed converter enables soft-switching for the main switch.

5. Experimental Results

After verifying the feasibility of the proposed converter using PSIM simulation software, the practical implementation of the high step-up soft switching converter was conducted using the TMS320F2809 [14] digital signal processor as the control core. The overall hardware circuit setup is shown in Figure 23, and the test platform is depicted in Figure 24. Figure 25 shows the waveforms of the trigger signal for the main switch S, input voltage, and output voltage of the high step-up soft-switching converter under full-load operation P=340W. From Figure 25, it can be observed that when the duty cycle of the main switch is 0.33 and the input voltage is 72V, the output voltage reaches 430V, achieving the expected voltage gain as shown in Table 1. Figure 26, Figure 27, Figure 28 and Figure 29 show the waveforms of the trigger signals, voltages, and currents for the main switch S and auxiliary switch S_r at load conditions of P=100W and P=340W, respectively. The experimental results demonstrate that the proposed high step-up ratio soft switching converter achieves ZVS for the main switch S under both light load (P=100W) and full load (P=340W) conditions. This helps reduce the switching loss, thereby improving the overall conversion efficiency. Additionally, the measured waveforms of the main switch S and auxiliary switch S_r match the simulated waveforms. Figure 30 shows a comparison of the efficiency between the proposed high step-up soft-switching and hard-switching converters, with loads ranging from P=100W to P=340W. The figure demonstrates that the conversion efficiency improves by 3-4% across different load levels.

6. Conclusion

This paper presents a high step-up soft switching converter topology. Through both simulation analysis and experimental results, it has been demonstrated that the converter achieves zero voltage switching (ZVS) for the main switch. By replacing the traditional energy storage inductor with a coupled inductor and adding a boost circuit, the overall voltage gain of the converter is enhanced, enabling it to achieve a higher voltage output. Additionally, a resonant branch is incorporated to ensure the main switch operates with ZVS, thereby reducing switching losses and switch stress, which in turn improves conversion efficiency. The proposed converter offers advantages such as ease of control via triggering signals, a simple circuit topology, and quantifiable component design parameters. Under different load conditions, the conversion efficiency improves by 3-4%, with a peak efficiency of approximately 93%. Thus, the converter demonstrates excellent conversion performance. This converter is promising for practical applications in photovoltaic module arrays, where it can be used for maximum power point tracking to enhance power generation efficiency.