A 32-GHz Eight-way Power Amplifier MMIC in 150nm GaN HEMT Technology

1. Introduction

Deep space exploration has been developed since the 1960s, and unmanned probes are exploring the far reaches of outer space and transferring data back to Earth. Early data transfers for deep space exploration were sufficient with a few kbps, but the demand for higher data rates has grown over time. It is essential to provide a large amount of communication capacity on spacecraft-to-Earth links to achieve this aim. As a result, deep space radio communication systems have moved toward higher frequencies, from L-band to S-band which was followed by S-band to X-band [1]. Moreover, the exponentially rising amount of investigated data and the need for higher transmission capacity are driving satellite communications to higher bands, such as those above the X band [2].

Millimeter-wave frequency band allows the deep-space satellite communication systems to reach high data transmission rates, with low communication delays. The Ka-band (27-40GHz) provides relatively wide bandwidth and high throughput. The ITU radio regulations assigned the downlink frequency ranges for the deep space exploration to X-band around 8.4GHz (8.4-8.45GHz) and Ka-band around 32GHz (31.8-32.3GHz). The Ka-band downlink deep space exploration systems can downlink data at a rate approximately six times higher than X-band [3]. However, because of the Earth’s troposphere, radio meteorological factors, convective rainfall, nonprecipitating clouds, stratiform rain events, atmospheric noise temperature, and other conditions, the attenuation might be much higher at Ka-band [4]. Since signal attenuation increases during propagation at high frequency, high-performance amplifiers in the system are vital in improving the link distance and the quality of communication [5].

To develop such a high-performance transceiver at millimeter-wave regime, high-power, low-cost, and highly reliable solutions for solid-state power amplifiers have been investigated in III-V compound semiconductor technologies like gallium arsenide (GaAs), gallium nitride (GaN). The III-V compound semiconductor technology has excellent transistor characteristics in the high-frequency range, and the semi-insulating substrate facilitates high-quality integrated passive components with low parasitic elements [6]. Specifically, GaN has a high breakdown voltage from a wide energy gap of 3.4eV, high carrier concentration, high electron mobility, and high saturation velocity. Because a high breakdown voltage transistor (HEMT) is available with a high current density in GaN technology, the PA can achieve a higher output power with improved power-added efficiency (PAE) at the millimeter-wave regime. This work presents a 32GHz high power amplifier (HPA) for a deep-space satellite communication system.

In Section 2, the power amplifier design is described. In the following, the simulation result and measurement result are presented in Section 3 and followed by a conclusion in Section 4.

2. 32-GHz High Power Amplifier Design

The designed 32GHz 8-way high power amplifier (HPA) was implemented in GaN/SiC HEMT technology with a gate length of 0.15-µm. The GaN heterostructure is grown on a SiC substrate and the source-coupled field plate structure provides an improved high breakdown voltage required for high-power RF applications. The technology supports the backside via-process to provide improved thermal conductance and reduced parasitic inductance. It is noteworthy that the backside via model from the process design kit (PDK) was tailored based on the measurements for a more accurate millimeter-wave PA design. The circuit and electromagnetic simulations have been performed using Keysight ADS.

The schematic of the 32 GHz HPA is presented in Figure 1. A cascaded two-stage common source (CS) amplifier topology is used to achieve more than 15dB gain at the operating frequency. To achieve an output power higher than 38dBm, an eight-way power combining network in the current mode was employed to effectively combine the power delivered from the parallel output stages. For each parallel HEMT, R=50 Ω is connected between each gate of the transistors to avoid any undesirable odd mode oscillation issue caused by the device asymmetry in the fabrication. As the HEMT devices are operating at 25V with high DC current, double metal layers were used for the bias line to minimize undesirable voltage drop.

2.1. Device Selection and Stabilization

In this work, an 150nm GaN HEMT technology was used. There are two types of HEMT models depending on the backside via layout configuration. One is the outside back via (OSV) HEMT and the other is the internal backside via (ISV) HEMT. With the same number of gate fingers and width, the OSV HEMT layout occupies a smaller area than that of the ISV HEMT. To minimize the chip area consumption, the OSV HEMT was used despite the fact that ISV HEMT has better self-heating performance with a large number of vias on the backside. The class-AB configuration was selected to improve the power efficiency, and the device was biased to the drain supply of 25V and the gate voltage of -1.55V. The size of the HEMT used at the first stage (H_1-4) was selected to be 6×50 μm and 8×50 μm HEMT was used for the final stage (H_5-12). Hence, the gate width periphery ratio is 3:8 to provide sufficient driving power margin between each driving stage.

In designing a robust high-power amplifier (HPA), stable operation of the HPA is essential. In this work, a parallel RC network at the gate was employed to improve stability, especially at a low-frequency region. Since the parallel RC network at the gate of each HEMT degrades the power gain and efficiency, it is critical to determine the proper values of the resistance and capacitance of the parallel RC networks for a high-performance HPA design. To guarantee unconditional stability from 100 MHz to 50 GHz, the parallel RC network was comprised of a 40 Ω resistor and 580fF capacitor for 6x50μm and a 50Ω resistor and 630fF capacitor for 8x50μm. Figure 2 shows the simulated maximum power gain (G_ma) and the μ- stability factor of the 6x50μm, and 8x50μm OSV layout devices. The G_ma at 32 GHz is reduced from 13.1dB to 12dB for 6x50μm, and from 14.4dB to 11.5dB for 8x50μm. Meanwhile, the device itself is unconditionally stable with μ > 1 from 100 MHz to 50GHz. Figure 2 presents the effects of the parallel RC network on the maximum power gain (G_ma) and μ-stability with and without the parallel RC network.

In combining multiple parallel HEMT devices, there is a possibility of odd-mode oscillations which can be caused by the electrical asymmetry from the differences in transconductance, biasing current, and nonlinear parasitic capacitors. It is noted that the odd-mode instability is not predictable from either μ or K-Δ stability factor from the small signal S-parameter simulations. To suppress unpredictable instability, resistors are placed between paralleled transistor branches [7].

Sufficient bypass capacitors are essential to suppress the noise, interferences, and RF leakages through the bias circuits. To improve the stability of the HPA, we added a de-Qing resistor in series with the bypass capacitor to suppress the ringing of the series RLC network from the parasitic routing inductance (L_para). It is important to select a proper series resistance since the de-Qing resistor trades off with the performance of the filtering capacitance. To keep the series RLC network in the over-damped oscillation, the series de-Qing resistor R_deQ should meet the condition given by

$R_{deQ}>2\sqrt{\frac{L_{para}}{C_{bypass}}}$,

where L_para is the routing inductance, and C_bypass is the bypass capacitance. It is noted that L_para should be minimized to keep R_deQ small enough not to degrade the filtering performance. The L_para are estimated from the ADS momentum simulation and the value of C_bypass is determined to meet the over-damped oscillation condition as a function of the de-Qing resistor in time domain simulations. The parameters of the series RLC network of the bypass capacitor with de-Qing resistors are presented in Figure 3.

A simplified block diagram of implemented stabilization circuits is illustrated in Figure 4.

2.2. Impedance Matching Network Design

In designing HPA, an ADS momentum simulator is used for optimizing transmission lines for accurate impedance matching at high frequencies. The design procedure involves careful EM simulation with backside vias formed through a thick SiC substrate to avoid frequency shift and impedance mismatch. In this work, an additional backside via inductance of 32 pH was included in series with the via model from PDK. To suppress undesired low-frequency signals throughout the matching network, the matching network was designed to have a high-pass property. The output matching network determines the power delivered to the load and the power added efficiency (PAE). Load-pull simulations were used to determine the optimum output-matching impedance. A load-pull simulation of 8x50μm OSV HEMT is presented in Figure 5. The power contour step size is 0.8 dB, and the PAE contour step size is 5%. The optimum impedance for the maximum power delivery was Z_{pout_opt} = 9.2 + j35.8 Ω and for the maximum PAE was Z_{PAE_opt} = 4.8 + j37.5 Ω. The input impedance of the output matching network from 28GHz to 32GHz is also shown in Figure 5. To achieve a high outpower level, the output matching network has an input impedance of Z_in = 12.7 + j36 Ω which is close to Z_{pout_opt} at 32GHz.

3. Simulation and Measurement Results

The proposed 32 GHz power amplifier was designed in 150nm GaN HEMT technology with a f_T of 34.5 GHz. Figure 6 shows a photograph of the implemented amplifier, which occupies an area of 4.3 mm × 4.5 mm including all the RF, ground, and DC pads. We measured the power amplifier’s S-parameters and large-signal performance. DC bias was supplied by ROHDE&SCHWARZ(R&S) HMP4040. S-parameters are measured with vector network analyzer R&S ZNB40. During large-signal measurements, the waveform generator Keysight 33250A generates a pulse-modulated signal with a 10% duty cycle and 1 ms pulse. The input signal was generated from a signal generator R&S SMA100B and for the measurement at a high input power level, a driving amplifier was used. To prevent returned signals from the reflection, an isolator Fairview Microwave FMIR1036 was used between the driving amplifier and the designed power amplifier. The output power was measured with a spectrum analyzer R&S FSV3044. The measurement setup for the S-parameter is illustrated in Figure 7. Figure 8 shows a setup for large signal measurement.

The measured small signal and large signal performances are presented in Figure 9. The gain at 32 GHz is 16.4 dB, and the 3-dB gain bandwidth was 3.5GHz between 31.3 and 34.8 GHz. The gate bias voltage was V_GG = -1.55 V and the drain supply voltage was V_DD = 25 V with a total drain current of 410mA. As shown in Figure 9a, the maximum gain was 16.9dB at 32.7GHz and the 3dB gain bandwidth was 3.5 GHz between 31.3 and 34.8 GHz. During the large signal test at 32 GHz under a 10% duty cycle and 1ms pulse, the P_sat was measured to be 38.3 dBm (6.76 W), and the power gain was 7.3 dB. The peak PAE was 19.3% and the output of 1 dB gain compression point (OP1dB) was around 28.5 dBm. Measurement results corresponded well with the simulations with small degradations at the high-frequency regime.

Table 1 summarizes the performance of this work in comparison with the recently reported Ka-band power amplifiers. The proposed design has the best small signal gain and 3dB gain bandwidth performance compare with other 2-stage designs in Ka-band.

4. Conclusions

We presented a 2-stage 8-way 32-GHz high power amplifier (HPA) in 150nm GaN HEMT technology for Ka-band deep space satellite communication systems. To achieve a high output power with minimum chip area, outside back via (OSV) type HEMT was used with the stabilizing circuitry using a parallel RC network at the gate, and bypass capacitor network with a de-Qing resistor. The values of stabilization resistors and capacitors are selected by minimizing gain degradation. With this configuration, the whole amplifier occupies 19.35 mm² including all the RF and DC pads. The measured amplifier demonstrated an output power of 38.3 dBm and 16. 4dB small signal gain at 32GHz with 3dB gain bandwidth of 3.5 GHz between 31.3 and 34.8 GHz which corresponds well with the simulation result.