A Fully Coaxial Tunable Broadband Power Equalizer for an X-band MPM

Microwave Power Modules (MPM) are high power devices largely used for radar applications. An MPM is a hybrid combination of a Solid-State Power Amplifier (SSPA) as a driver and a Traveling wave tube amplifier (TWT) as the final high power stage. Power equalizer is often used to flat the TWT output power gain owing to its big gain fluctuations in the operating frequency range. This paper deals with an optimal design and development of a fully-coaxial easy-tunable power equalizer for an X-band MPM. The structure utilizes a coaxial waveguide as the main transmission line and a coaxial cavity loaded with absorbing material as the resonant unit. A sensitivity analysis of the coaxial cavity length, the thickness of the absorbing disc and inserted depth of the probe on the attenuation amplitude and resonant frequency is carried out. The transmission characteristic curves and the voltage standing wave ratio curves show a good compromise in terms of transferred power and adaptive matching impedance at the input and output ports. The measured results show that the equalization curve meets the requirements well and prove that this structure is practical and effective.


INTRODUCTION
Wide-band high-power sources operating at microwave and millimeter wave necessitates more efficient devices [1][2] [3]. Microwave Power Module (MPM), a hybrid combination of a Solid-State Power Amplifier (SSPA) as a driver and a Travelling wave tube amplifier (TWT) as the final high power stage, is widely used in many potential system applications, such as radar and electronic warfare. The gain equalizer is one of the important parts in MPM, which is used to flatten the output gain of traveling wave tube [4][5] [6].
Due to the standardization necessary to keep production costs low, the manufacturer of these equalizers generally designs and produces his own network not considering the problems of the circuits connected to it. As a result, equalizers must be mounted separately, requiring additional A Fully Coaxial Tunable Broadband Power Equalizer for an X-band MPM cables and connectors, which results in greater or lesser compensation than desired.
Since the requests of the users of such devices have become rather demanding, as regards what constitutes acceptable operating characteristics or parameters when such devices are used in apparatuses having ever more rigorous and involved design criteria, the redesign of the device for producing the required output characteristic is prevented by the efficiency of this choice.
To get the advantages of small size, lightness, simple structure and ease of processing, the microstrip circuit is the most used for the realization of the equalizer [7][8] [9]. In this case, however, the costs of redesigning the device, depending on the requests of the user, limit its use.
To have the advantages of a high factor of merit, low transmission losses, and a tuning capacity, a coaxial equalizer must be realized [10] [11].
However, to be economic, such an equalizer should not show the usual high-cost increases usually associated with device miniaturization. Of course, if a smaller, lighter, and lower-cost microwave equalizer were available with a low Voltage Standing Wave Ratio (VSWR) compared to the normal class of microwave equalizers, its contribution to the art would be evident.
In this paper, the design and realization of a fully coaxial tunable broadband equalizer for an Xband MPM, are reported [12].

II. STRUCTURE AND ANALYSIS
The microwave amplitude equalizer design consists of several substructures formed by absorbent coaxial resonant cavities. Many factors, such as the length of the cavity, the size of the cavity and the probe, the materials, the position of the adjustable metal micrometric screw, the attenuation screw, the number of resonators, etc., have been taken into consideration in the design [10]. The schematic of the fully coaxial tunable equalizer is reported in Fig. 1. L1 is the cavity length, L2 the insertion depth of the probe, and D the thickness of the absorbing disc. .

c. Length of the resonator
The length l of the coaxial resonator l is determined from the following relation: where λo is the resonance wavelength.
Since the resonance wavelength is equal to multiples of λ /4, the resonator is called a resonant cavity at λ /4.
With reference to equation (6) and considering that the extremes of the operating band are 8 and 12 GHz, it is possible to obtain the length of the cavity which in the worst case will be 9.5mm.

e. Absorbing disc thickness
The thickness of the absorbing disc D is directly connected to the thickness of the resonator resistance. So, an increasing of the absorbing disc thickness will increase the resistance R and according to eq. (1) a decreasing in Q will occur.
By keeping the length of the cavity and the insertion depth of the probe constant, the transmission coefficient S21 and the input VSWR curves vs frequency with the thickness of the absorption cylinder as a parameter have been drawn and shown in Fig.4. It can be observed that the thicker the absorption cylinder, the lower the resonance frequency and the attenuation, with a lower input VSWR. A disk thickness of 0.5 mm as a good compromise between S21 and VSWR has been chosen. III.

SIMULATIONS
In Fig.5, the schematic design of the proposed equalizer using 4 coaxial cavity resonators in cascade is reported. The equalizer has been designed to match the amplifier with the power attenuation curve at each working frequency.  In Fig. 7, the input and output VSWR of the designed equalizer are shown. The input VSWR has a peak value of about 2.2, which corresponds to a reflected power of about 13%. To keep the VSWR low, an absorbent disc that is not too thin or a probe that is not too coupled have to be use. In both cases this leads to a decrease in attenuation. Since, the desired attenuation is lower than that simulated, then this trade-off can easily be solved. Starting from the structure shown in Fig.5 and with the help of HFSS by Ansoft, it is possible to model and simulate the single resonator disconnecting all the others.
The simulations of the S21 and VSWR curves have been carried out as follows: -L2 and D constant and L1 as a parameter, shown in Fig. 8; -L1 and D constant and L2 as a parameter, shown in Fig. 9.
The curves show a good compromise in terms of transferred power and adaptive matching impedance at the input and output ports.

IV. MECHANICAL STRUCTURE
The external design of the device is shown in Fig. 10. It has a height, width, and depth of 15 mm, 9 mm, and 40 mm respectively. It is made by perforating an aluminum parallelepiped and inserting in it the tuning screws, the absorption disc, and the main transmission line interfaced with the outside through the glass bead.  The sensitivity analysis of the insertion of the glass bead into the structure and of the distance of the main line between the input and output port, carried out with HFSS, is reported in Fig.11. Figure 11 a) shows the more you insert the glass bead in the structure, the less the attenuation will be. Figure 11 b) shows that the more the distance of the main line between the input and output port the more the mismatch will be. In this paper, the design of a fully coaxial broadband tunable microwave equalizer was presented.
The structure has the advantage of large power capacity, wide operating frequency band, convenient tune and good transmission performance. The equalizer operating at X-band meets the design requirements well to make flat the output characteristic of an MPM for radar applications. VI.